The present invention relates generally to transmitter methods and apparatus and, more particularly, to a transmitter method and apparatus wherein energy in a portion of a spectrum on a first side of a predetermined frequency is transferred to a second side of the predetermined frequency in such a manner that at least some of the transferred energy results in a reduction of transmitted bandwidth and receivers responsive to the transmitter respond to the transferred energy as if it were on the first side of the predetermined frequency.
The problem of frequency spectrum crowding has become so severe that government allocated spectra for different types of transmission results, in some cases, in overlapping spectra for different transmitters. For example, the United States Federal Communications Commission (FCC) in establishing a table of channel allocations for the Advanced Television Standards Committee (ATSC), eight level vestigial sideband (8VSB) digital television (DTV) transmitters has created severe technical problems for television channel 14 (470-476 MHz). There are terrestrial mobile radio transmitters which the FCC has licensed at 469.975 MHz, i.e., at a frequency allocation very close to the lower band edge of channel 14. It is extremely difficult to protect the low power 469.975 MHz radio emission from the high power DTV signal having energy at 470 MHz, only 0.025 MHz away from the 469.975 MHz emission. This is particularly the case since the FCC has required the 470 MHz band edge to be down only 36 dB from the mid-band amplitude. If prior techniques were used, the channel 14 allocation would be useless in most cases because it would overpower the 469.975 MHz emission, a situation the FCC will not permit.
Another situation that is problematic is the so-called xe2x80x9cN+1xe2x80x9d allocation. In the xe2x80x9cN+1xe2x80x9d allocation, a National Television Systems Committee (NTSC) licensee is assigned a DTV channel that is the next channel up from the licensee""s NTSC channel. In this situation, the NTSC aural carrier has significant sidebands that extend to within about 75 kHz of the NTSC channel edge, and therefore to within about 75 kHz of the DTV signal. This creates problems for stations that wish to use frequency selective combiners with separate NTSC and DTV transmitters. Yet another problem is the radio telescope allocation at channel 37 in the UHF DTV band (608-614 MHz). DTV stations on channel 36 or 38 may interfere with radio telescope operations on channel 37.
A typical prior art digital television transmitter adapted to transmit signals containing information indicative of digitally encoded video and aural signals for deriving an ATSC A/53 standard signal is illustrated in FIG. 1 as including multi-bit digital baseband television signal source 10 which drives the cascaded combination of data randomizer 11, Reed-Solomon encoder 12, data interleaver 14, trellis encoder 16 and multiplexer 18. In the 8VSB system, the signals derived from source 10, randomizer 11, encoders 12 and 16, as well as interleaver 14 and multiplexer 18 include three parallel bits having a symbol rate (i.e., sampling frequency) of             1.539      xc3x97              10        9              143    ,
i.e., the encoded television signal that source 10 derives is sampled 10,762,237.76 times per second. Because each symbol includes two or three bits, the bit rate is substantially higher than the symbol rate. The three parallel bits represent 8 amplitude levels of the encoded television signal.
Multiplexer 18, in addition to being responsive to the output of trellis encoder 18, responds to segment synchronizing source 20 and field synchronizing source 22 to derive an output having the same number of bits as applied to the multiplexer by encoder 16. Multiplexer 14 supplies a multi-bit output signal to pilot inserter 24 which inserts a constant bit pattern representing a DC amplitude equal to 0.625 of the weight of a single bit in the incoming 3 bit pattern. Pilot inserter 24 derives a multi-bit output signal which drives pre-equalizer filter 26.
Pilot inserter 24 derives a multi-bit output signal which drives pre-equalizer filter 26. Pre-equalizer filter 26 supplies a multi-bit intermediate frequency (IF) signal to vestigial sideband modulator or generator 28. Generator 28 feeds a multi-bit digital IF signal to digital to analog converter 30, which supplies an analog IF signal to frequency up converter 32. Converter 32 is a frequency synthesizer that heterodynes the IF output frequency of converter 30 to a radio frequency (RF) carrier frequency. Up converter 32 also inverts the IF spectrum digital to analog converter 30 derives so up converter 32 converts (1) the lowest frequencies in the IF spectrum into the highest frequencies in the RF spectrum and (2) the highest frequencies in the IF spectrum to the lowest frequencies in the RF spectrum. RF up converter 32 applies the modulated carrier frequency signal to antenna 34 via power amplifier 36.
The output signal of digital to analog converter 30 includes orthogonal I and Q channels or components. At predetermined time intervals, after the receiver""s root raised cosine filtering, the I channel has one of multiple levels, corresponding to the number of amplitude levels in the 3 bit signal source 10 derives. The Q channel contains no independent information, but causes part of the unwanted lower sideband appearing at the output of up converter 32 to be reduced substantially to zero amplitude. The unwanted lower sideband is removed by circuitry included in vestigial sideband generator 28 and up converter 32 does not reintroduce it. Because up converter 32 xe2x80x9cflipsxe2x80x9d (i.e., inverts) the IF spectrum that digital to analog converter 30 derives, the upper sideband RF output of converter 30 is reduced substantially to zero.
To enable digital to analog converter 30 to produce the desired vestigial sideband signal, vestigial sideband modulator or generator 28 derives the baseband vestigial sideband spectrum illustrated in FIG. 2. The spectrum of FIG. 2 has a 6 MHz bandwidth and includes the 309.44056 kHz pilot carrier frequency that pilot inserter 24 derives, as well as a vestigial sideband that extends 309.4405594 kHz below the pilot carrier frequency, to the left of the carrier as illustrated in FIG. 2. The response curve of FIG. 2 has a flat portion 38 that extends throughout the vast majority of the transmitter 6.0 MHz bandwidth. The response curve has monotonic root raised cosine (RRC) transitions 40 and 42 at its upper and lower band edges. Transitions 40 and 42 are symmetrical, each having a frequency extent of about 619 kHz (i.e., twice the pilot carrier frequency) between the edges of the transmitter bandwidth and the corner frequencies where the transitions end and flat portion 38 begins. Receivers responsive to transmitters of the type illustrated in FIG. 1 include a filter with the same response as FIG. 2.
The amplitude response of each of transitions 40 and 42 at any frequency (f) from the mid-point frequency (ft) of each transition is:                               Rrc          ⁡                      (            f            )                          =                  sqrt          ⁡                      (                                          1                2                            [                              1                +                                  cos                  ⁢                                      xe2x80x83                                    ⁢                                      π                    ⁡                                          (                                                                        f                          -                                                      f                            t                                                                                                    2                          ⁢                                                      xe2x80x83                                                    ⁢                                                      f                            t                                                                                              )                                                                                  )                        )                                              (        1        )            
Vestigial sideband modulator 28 for deriving the ATSC A/53 standard has generally used a Hilbert filter or phasing method. My co-pending commonly assigned application, Ser. No. 09/239,668, filed Jan. 29, 1999, entitled xe2x80x9cVestigial Sideband Generator Particularly for Digital Television,xe2x80x9d discloses a modified Weaver modulator as the device for generating the vestigial sideband spectrum that modulator 28 derives.
The ATSC standard 8VSB transmission system, like many other digital transmission systems, includes a certain amount of xe2x80x9cexcess bandwidth.xe2x80x9d Because the 8VSB transmitter transmits symbols at a rate of 10.76223776 . . . million per second, the transmitter requires a minimum theoretical bandwidth of 5.381118881 . . . MHz (half the symbol rate). Although this is a theoretical minimum, it is physically impossible to build a transmitter that only uses the minimum bandwidth. As a practical matter, it is necessary to have a certain amount of additional (xe2x80x9cexcessxe2x80x9d) bandwidth for filter transition bands. Consequently, the 8VSB system has about 11.5% excess bandwidth. In other words, the 6 MHz channel width divided by 5.381118881 . . . MHz is approximately 11.5% greater than unity. To achieve an overall flat amplitude response between a DTV transmitter and a DTV receiver, the ATSC standard requires the shape of filtering in transition bands of the edges of the spectrum to have the root raised cosine (RRC) response of Equation (1). Because the RRC filter response is applied twice, at the transmitter and at the receiver, the 8VSB signal magnitude response is squared, providing an overall raised cosine shape. When the receiver demodulates the signal it receives from the transmitter, the I channel of the 8VSB signal has a flat amplitude response.
In the Hilbert filter method, vestigial sideband modulator 28 generates a double sideband signal that is filtered to produce a vestigial sideband signal at an IF of about 10 MHz. Sidebands extend equally around the 10 MHz IF in accordance with:
0.5Fsym+Fpilot=6MHzxe2x88x92Fpilotxe2x80x83xe2x80x83(2)
where Fsym is the 10.76223776 . . . MHz symbol clock frequency of the bits source 10 derives in accordance with the ATSC A/53 standard, and       F    pilot    =                    59        3            ·              FH        NTSC              =                  59        3            ·                        4.5          ⁢                      xe2x80x83                    ⁢          MHz                286            
where FHNTSC is the NTSC horizontal line frequency.
Based on the foregoing, a vestigial sideband modulator 28 including a Hilbert transform has a double sideband modulator with sidebands that extend xc2x15.690559441 . . . MHz on either side of the about 10 MHz carrier. A convenient sampling frequency is four times the 10.76223776 . . . MHz symbol clock rate, i.e., 43.04895105 . . . MHz.
The Hilbert transform modulator uses a phasing method that partially cancels the unwanted sideband of the double sideband signal with the RRC transition. Obtaining a proper root-raised cosine response for vestigial sideband shaping at the 43.04895105 . . . MHz sampling rate of the Hilbert transform requires a finite impulse response (FIR) filter having about 2048 filter coefficients. Implementation of such a filter is difficult.
The Hilbert transform can easily generate a vestigial sideband signal such that DC is 6 dB down with respect to the sidebands. This is because the response of any Hilbert transform approximation is always zero at DC. With only one of the I and Q modulators included in such a vestigial Hilbert transform sideband generator contributing at DC, the vector sum of the outputs of the two modulators drops in half at DC relative to the vector sum at a frequency where both the I and Q channels contribute to the generator output. However, in the ATSC A/53 standard, the requirement for the root-raised cosine response places the DC output at xe2x88x923 dB instead of xe2x88x926 dB. Therefore, the Hilbert transform method of vestigial sideband digital television modulation requires a low frequency equalizer to produce a +3 dB xe2x80x9cshelfxe2x80x9d at DC and the low frequency portions of the baseband spectrum.
To achieve the ATSC A/53 standard, the Hilbert transform vestigial sideband generator has a linear phase requirement. Consequently, equalizer filter 26 is generally implemented as a finite impulse response filter having a large number of coefficients. Further, a xe2x88x923 dB requirement exists at the Nyquist frequency of the symbols, i.e., half the symbol frequency, with certain modifications. Hence, equalizer 26 must include a high frequency portion operating at a sampling frequency higher than twice the symbol rate to avoid aliasing, i.e., insertion of information at frequencies that do not exist in the sample frequency due to sampling at a frequency less than twice the highest frequency component being filtered. In this case, the highest frequency being filtered is 5.690559441 . . . MHz, which is more than half the symbol rate. This is another reason why the Hilbert transform method of producing a vestigial sideband signal with root-raised cosine sideband shaping is also quite difficult to implement.
Because power amplifier 34 has a non-linear amplitude response, nonlinear equalizer 26 must apply a substantial non-linear correction to the signal applied to it. Because of the possibility of aliasing and spectral folding through zero frequency, the amount of nonlinear correction which can be applied at 10 MHz is limited, resulting in distortion in the transmitted signal.
My co-pending application is based on my realization that the non-linear correction can be more effectively provided by using the modified Weaver modulator in a way that substantially reduces distortion in the transmitted signal by employing an IF digital signal having a frequency approximately twice the approximately 10 MHz frequency of the prior art digital IF. The arrangement I previously invented and disclosed in my co-pending application enables the digital IF signal to have a frequency of approximately or exactly 21.5 MHz.
The modified Weaver vestigial sideband modulator includes an oscillator with a folding frequency and lowpass filters with cut off frequencies such that xe2x80x9cnegativexe2x80x9d frequencies the modulating baseband signal pilot inserter 24 derives appear in a signal derived from an adder as a vestige of the opposite sideband, i.e., the positive frequencies of the modulating baseband signal. The folding frequency and the cutoff frequency of the lowpass filters are selected such that a desired amount of the opposite sideband is eliminated but a certain portion of it is passed. Preferably, vestigial sideband modulation is produced by using a Weaver modulator having a reduced folding frequency and lowpass filters having increased cutoff frequencies.
It is, accordingly, an object of the present invention to provide a new and improved method of and apparatus for reducing the bandwidth of a transmitted signal without affecting a conventional receiver responsive to the transmitted signal.
Another object of the invention is to provide a new and improved method of and apparatus for generating a vestigial sideband signal, particularly a digital vestigial sideband IF signal.
An additional object of the invention is to provide a new and improved digital television transmitter apparatus and method wherein a tendency for the transmitted signal in some channels to interfere with signals having a neighboring spectrum is overcome.
A further object of the invention is to provide a new and improved digital television transmitter apparatus and method wherein a tendency for the transmitted signal in some channels to interfere with signals derived by neighboring low power transmitters is avoided.
An added object of the invention is to provide a new and improved 8VSB transmitter employing digital filtering techniques for modifying the 8VSB spectrum in such a manner as to avoid interference with neighboring transmitters, wherein the 8VSB transmitter includes passive analog channel output filters and combiners and standard DTV receivers respond to the modified 8VSB spectrum to provide a television signal that is not degraded.
According to one aspect of the invention, a method of preventing substantial interference between portions of first and second spectra that tend to overlap is provided. The overlapping portion of the first spectrum is in a band of frequencies on a first side of a predetermined frequency of the first spectrum. The predetermined frequency is outside the second spectrum and within the first spectrum. The method includes the steps of modifying the first spectrum prior to transmission thereof by transferring energy in frequencies on the first side of the predetermined frequency of the first spectrum to frequencies on the second side of the predetermined frequency so that all energy between a first frequency at a band edge of the first spectrum and a second frequency between the first frequency and the predetermined frequency is reduced substantially to zero. The frequencies on the second side are within the first spectrum. The transfer is such that a receiver designed to respond to the first spectrum causes the transferred energy to be shifted from the second side of the predetermined frequency to the first side of the predetermined frequency. The amount of energy transferred is such that there is no substantial interference between the transmitted first and second spectra.
The receiver includes a bandpass filter with a transition having a predetermined amplitude versus frequency response between a frequency at one band edge of the bandpass filter and a mid-range portion of the bandpass filter response. The predetermined frequency is displaced toward the frequency at said one band edge and from a corner frequency between the mid-range portion and the transition. The energy transfer is preferably such that (1) all energy between the first and second frequencies is transferred on a frequency to frequency basis to frequencies that are on the second side of the predetermined frequency and are equally displaced from the predetermined frequency, and (2) some energy between the second frequency and the predetermined frequency is transferred, on a frequency to frequency basis, to frequencies that are on the second side of the predetermined frequency and are equally displaced from the predetermined frequency. The amount of the energy transferred from between the second frequency and the predetermined frequency is determined by the relative amplitudes of (a) an amplitude versus frequency weighting function that extends over a band of frequencies between the second frequency and the predetermined frequency and (b) the amplitude versus frequency response of the receiver transition.
The invention is particularly useful for situations wherein the first spectrum is a DTV vestigial sideband spectrum and the receiver bandpass filter has a flat mid-range amplitude versus frequency response and a root raised cosine transition between the corner frequency and the first frequency. In such a case, the predetermined frequency at the low frequency band edge is a baseband pilot carrier frequency and at the high frequency band edge is a Nyquist sampling frequency.
In one embodiment, the modified spectrum has a symmetrical amplitude versus frequency response with respect to its center frequency. In another embodiment, the modified spectrum has an amplitude versus frequency response at its high frequency band edge including the transferred energy and an amplitude versus frequency response at its low frequency band edge that is substantially the same as the receiver transition. In a further embodiment, the modified spectrum has an amplitude versus frequency response at its low frequency band edge including the transferred energy and an amplitude versus frequency response at its high frequency band edge that is substantially the same as the receiver transition.
Preferably, the energy transfer includes lowpass filtering a signal resulting from a baseband signal containing information to be transmitted. The lowpass filtering includes multiplying the signal resulting from the baseband signal by (a) a constant amplitude from DC to another frequency where the transition begins, (b) amplitudes greater than the constant amplitude for frequencies between the another frequency and a further frequency, (c) finite amplitudes less than the constant amplitude for frequencies greater than the further frequency and (d) zero for frequencies that are displaced from the another frequency by the same amount that said frequency at the band edge of the bandpass filter is displaced from the corner frequency.
In one embodiment, the energy transfer includes multiplying a baseband signal containing information to be transmitted by sine and cosine representations to derive a pair of product signals. The sine and cosine representations are at a folding frequency. Replicas of the product signals are lowpass filtered as described supra. The signals resulting from the lowpass filtering are combined to derive the reduced bandwidth spectrum.
In one embodiment, the replicas on which the lowpass filtering steps are performed result directly from the multiplying operations to cause the spectrum resulting from the combining step to be symmetrical with respect to a frequency at the center of the first spectrum.
According to another embodiment, second lowpass filtering steps are performed on replicas of the product signals. The second lowpass filtering steps are performed on the product signal replicas before the first mentioned lowpass filtering operation is performed. The second lowpass filtering operations include multiplying the replicas of the product signal by (a) a constant amplitude from DC to the another frequency, and (b) the receiver transition amplitude versus frequency response for frequencies between the another frequency and a frequency removed from the another frequency by an amount equal to the displacement of the band edge of the receiver bandpass filter from the corner frequency. The spectra resulting from the second lowpass filtering steps are shifted to cause the spectrum resulting from the combining step to be asymmetrical with respect to a frequency at the center of the first spectrum. The asymmetrical spectrum has the same amplitude versus frequency response at a first band edge as the receiver bandpass filter and has an amplitude versus frequency response at a second band edge determined by the first mentioned lowpass filtering operation.
In another embodiment, the signal resulting from the first mentioned lowpass filtering step is multiplied by sine and cosine representations to derive a pair of product signals. The sine and cosine representations are at a folding frequency. Second lowpass filtering operations are performed on replicas of the product signals. The second lowpass filtering operations are performed on the product signal replicas before the first mentioned lowpass filtering operation is performed. The second lowpass filtering operations multiply the replicas of the product signal by (a) a constant amplitude from DC to the another frequency, and (b) the receiver transition amplitude versus frequency response for frequencies between the another frequency and a frequency removed from the another frequency by an amount equal to the displacement of the band edge of the receiver bandpass filter from the corner frequency.
The signals resulting from the second lowpass filtering step are preferably multiplied, according to one embodiment, by cosine and sine representations of an IF to derive second product signals. The second product signals are combined to derive an asymmetrical spectrum that has the amplitude versus frequency response of the receiver transitions at its low frequency edge and the amplitude versus frequency response resulting from the first mentioned lowpass filter operation at its high frequency edge.
According to a further embodiment, I and Q channel Hilbert transform operations are performed on a signal resulting from the baseband signal so that signals resulting from the I and Q channel transform operations have the first mentioned lowpass filtering operations applied to them. Replicas of the signals resulting from the I and Q channel Hilbert transform operations are combined to derive the first spectrum. The combining step preferably includes multiplying the signals resulting from the I and Q channel Hilbert transform operations with representations of cosine and sine waves at an IF to form product signals that are added.
A further aspect of the invention relates to a transmitter for a first spectrum having a portion that tends to overlap in an interfering manner with a neighboring spectrum. The overlapping portion is in a band of frequencies on a first side of a predetermined frequency that is outside the second spectrum and within the first spectrum. The transmitter includes a filter for modifying the first spectrum prior to transmission. The filter is arranged for transferring energy in frequencies on the first side of the predetermined frequency of the first spectrum to frequencies on a second side of the predetermined frequency. The energy transfer causes substantial elimination from the first spectrum of all energy between a band edge of the first spectrum and another frequency between the band edge and the predetermined frequency. The frequencies on the second side are within the first spectrum. The transfer is such that receivers designed to respond to the first spectrum cause the transferred energy to be shifted from the second side of the predetermined frequency to the first side of the predetermined frequency. An output device is connected to be responsive to the modified spectrum.
The filter preferably includes a digital vestigial sideband modulator connected to be responsive to a digital baseband television signal source and a carrier pilot inserted on the digital baseband television signal source.
In one embodiment, the digital vestigial sideband modulator includes a digital filter including a Hilbert transform combined with a lowpass filter having an amplitude versus frequency response for causing the energy transfer.
In a second embodiment, the vestigial sideband modulator includes a digital folding frequency source for deriving orthogonal components at the predetermined frequency. A digital multiplier arrangement responds to the orthogonal components and the digital signal source for deriving a pair of orthogonal product digital signal components. A digital lowpass filter arrangement responds to the pair of orthogonal product digital signal components. The orthogonal digital signal components passed by the digital lowpass filter arrangement are combined. The digital lowpass filter arrangement has a response and the digital signal components passed by the digital lowpass filter are combined so the energy transfer occurs and there is no substantial energy between the another frequency and the band edge of the first spectrum.
A further aspect of the invention relates to a memory for controlling an electromagnetic wave transmitter that is arranged for transmitting a signal to a receiver having a predetermined response. The memory stores signals for preventing substantial interference between portions of first and second spectra that tend to overlap. The overlapping portion of the first spectrum is in a band of frequencies on a first side of a predetermined frequency of the first spectrum. The predetermined frequency is outside the second spectrum and within the first spectrum. The stored signals cause a modification of the first spectrum prior to transmission thereof by transferring energy in frequencies on the first side of the predetermined frequency of the first spectrum to frequencies on a second side of the predetermined frequency so that all energy between a first frequency at the band edge and a second frequency between the first frequency and the predetermined frequency is reduced substantially to zero. The frequencies on the second side are within the first spectrum. The transfer is such that a receiver designed to respond to the first spectrum causes the transferred energy to be shifted from the second side of the predetermined frequency to the first side of the predetermined frequency.